Predictive feedback compensation for PWM switching amplifiers

ABSTRACT

Methods and systems are disclosed for predictive feedback compensation (PFC) circuitry for suppressing distortions caused by supply voltage variations and output amplitude switching non-idealities in pulse width modulated (PWM) switching amplifiers by pre-compensating the PWM input based upon the supply voltage or output pulse amplitude. Output amplitude errors associated with previous PWM output signals are used to predict output amplitude errors expected for future PWM output signals. These predicted output amplitude errors are then used to adjust the pulse widths for the future PWM output signals. Traditional feedback techniques can also be used in conjunction with the predictive feedback compensation (PFC) circuitry.

RELATED APPLICATIONS

This application claims priority to the following co-pending provisionalapplication: Provisional Application Ser. No. 61/128,412, filed on May21, 2008, and entitled “PREDICTIVE FEEDBACK EQUALIZATION FOR PWMSWITCHING AMPLIFIERS,” which is hereby incorporated by reference in itsentirety. The application is also related in subject matter to thefollowing concurrently filed application Ser. No. ______, entitled“CLOSED LOOP TIMING FEEDBACK FOR PWM SWITCHING AMPLIFIERS USINGPREDICTIVE FEEDBACK COMPENSATION” by John M. Khoury et al., which ishereby incorporated by reference in its entirety.

TECHNICAL FIELD OF THE INVENTION

This invention relates to Class D amplifiers and, more particularly, tosuppressing distortion and noise caused by power supply variations andoutput amplitude switching non-idealities in Class D amplifiers.

BACKGROUND

Performance of Class D switching amplifiers is susceptible todegradations from power supply variations and switching non-idealities.Power supply variations constitute a significant source of error sinceat full scale modulation the power supply rejection (PSR) is essentially0 dB. While analog feedback techniques have been successfully employedwith analog PWM (pulse width modulation) amplifiers to mitigate thesedegradations, applying feedback to digital PWM amplifiers is problematicbecause of incompatible domains and processing latencies.

One prior approach for mitigating these degradations employs open loopdigital pre-compensation as depicted in FIG. 1A (Prior Art). Theswitching amplifier embodiment 100 receives digital PCM (pulse codemodulated) signals and processes them with volume (VOL) control block102. The output of the volume (VOL) control block 102 is provided to thePWM (pulse width modulated) controller 104. PWM controller (PWM) 104outputs PWM signals to driver 106. The driver 106 provides the PWMoutput signals (PWM_(OUT)) for the Class D switching amplifier. To helpadjust for errors in the PWM output signals (PWM_(OUT)) due to voltagesupply variations, this prior solution feeds the supply voltage (Vp) forthe driver 106 to an analog-to-digital converter (ADC) 108 and then tofilter 110 to provide a feedback signal to the volume (VOL) controlblock. The gain applied by the volume (VOL) control block 102 to theincoming PCM signals is then adjusted based upon the feedback signalreceived from the filter 110. This prior approach, therefore, attemptsto compensate for amplitude errors in the output signals caused byvariations in the voltage supply (Vp) through voltage supply (Vp)feedback signals that adjust the amplitude of the incoming PCM signals.

Another prior approach employs closed loop feedback of the PWM pulsearea as depicted in FIG. 1B (Prior Art). The switching amplifierembodiment 150 includes a PWM controller (PWM) 104 that receives the PCMsignals and outputs PWM signals to a pulse edge error correction (PEDEC)block 152. The output signals from PEDEC block 152, which are edgecorrected PWM signals, are provided to driver 106. The driver 106provides the PWM output signals (PWM_(OUT)) for the Class D switchingamplifier. To help adjust for errors in PWM output signals (PWM_(OUT)),this prior solution sends the PWM output signal (PWM_(OUT)) as afeedback signal to an error processing block 154. The error processingblock 154 also receives the PWM input signals from PWM controller 104 asreference signals. The error processing block 154 then outputs edgeerror correction signals to the PEDEC block 152. The PEDEC block 152uses these edge error correction signals to adjust the edges of the PWMinput signals so that the PWM output signals 156 from the PEDEC block152 are edge corrected PWM signals. This prior approach attempts tocompensate for PWM pulse area errors in the output signals by comparingthe pulse area of the PWM output signal with that of the PWM inputsignal and then adjusting the edges of the PWM signals to compensate forthe area differences.

While these approaches have been employed to mitigate non-ideal effectsof digital PWM amplifiers, solutions are lacking that improve theintrinsic power supply rejection, distortion, and damping performance ofopen loop switching amplifiers.

SUMMARY OF THE INVENTION

Methods and systems are disclosed for predictive feedback compensation(PFC) circuitry for suppressing distortions caused by supply voltagevariations and output amplitude switching non-idealities in pulse widthmodulated (PWM) switching amplifiers by pre-compensating the PWM inputbased upon the supply voltage or output pulse amplitude. Outputamplitude errors associated with previous PWM output signals are used topredict output amplitude errors expected for future PWM output signals.These predicted output amplitude errors are then used to adjust thepulse widths for the future PWM output signals. Traditional feedbacktechniques can also be used in conjunction with the predictive feedbackcompensation (PFC) circuitry. As described below, other features andvariations can be implemented and related methods and systems can beutilized, as well.

In one embodiment, a method for correcting output amplitude errors inswitching amplifiers driven by pulse width modulated (PWM) signals isdisclosed. This method includes receiving a pulse width modulated (PWM)input signal having an input pulse width, predicting an output pulseamplitude error for the PWM input signal based on a prior PWM outputsignal, pre-compensating the input pulse width for the PWM input signalwith a width adjustment based upon a ratio of the predicted output pulseamplitude error to an output pulse amplitude weighted by a pulse width,and outputting a PWM output signal through a switching amplifier, thePWM output signal having a pulse width based upon the pre-compensatedpulse width for the PWM input signal. In one further embodiment, thepre-compensating step can include pre-compensating the input pulse widthfor the PWM input signal with a width adjustment based upon a ratio ofthe predicted output pulse amplitude error to an output pulse amplitudetotal value weighted by the input pulse width for the PWM input signalto produce the pre-compensated pulse width for the PWM input signal. Inanother further embodiment, the pre-compensating step can includepre-compensating the input pulse width for the PWM input signal with awidth adjustment based upon a ratio of the predicted output pulseamplitude error to an output pulse amplitude desired value weighted by apre-compensated pulse width for a prior PWM input signal to produce thepre-compensated pulse width for the PWM input signal.

Further, the pre-compensating steps can be implemented by adjusting thepulse width of only one edge of the PWM input signal or by adjusting thepulse width of both edges of the PWM input signal (e.g., symmetric ornon-symmetric). Still further, two PWM input signals can be receivedsuch that signal information resides in a difference between the twosignals, and pulse widths for each of the two PWM input signals can bepre-compensated prior to being output as two PWM output signals. Inaddition, output pulse amplitude error can be associated with a singleprior PWM output signal or the an output pulse amplitude errorassociated with a plurality of prior PWM output signals.

Still further, the predicting step can be implemented by measuring avarying or alternating current (AC) component of a supply voltage topredict the output pulse amplitude error for the PWM input signal basedon a prior PWM output signal. And this supply voltage measurement can beimplemented by comparing a total supply voltage to a reference voltagerepresenting a desired output pulse amplitude to measure the varying oralternating current (AC) component of the supply voltage. In a furtherembodiment, the predicting step can be implemented by measuring avarying or alternating current (AC) component of an output pulseamplitude for the PWM output signal to predict the output pulseamplitude error for the PWM input signal based on a prior PWM outputsignal. And this output amplitude measurement can be implemented bycomparing an output pulse amplitude total value to a reference voltagerepresenting a desired output pulse amplitude to measure the varying oralternating current (AC) component of the output pulse amplitude for thePWM output signal.

In another embodiment, a digital switching amplifier having output pulseamplitude error correction is disclosed. The digital switching amplifiercan include amplitude error prediction circuitry, width adjustmentcircuitry and a switching amplifier driver circuitry. The amplitudeerror prediction circuitry can be configured to sense a voltagerepresenting the output pulse amplitude for a PWM output signal, todetermine a predicted output pulse amplitude error for a PWM inputsignal using the sensed voltage, and to output a predictive errorcorrection signal proportional to a ratio of the predicted output pulseamplitude error to an output pulse amplitude weighted by a pulse width.The width adjustment circuitry can be coupled to receive the predictiveerror correction signal and a PWM input signal having a pulse width andcan be configured to output a pre-compensated PWM signal with apre-compensated pulse width representing a width adjustment based uponthe predictive error correction signal. And the switching amplifierdriver circuitry can be configured to receive the pre-compensated PWMsignal and to drive a PWM output signal.

In one further embodiment, the predictive error correction signal can bebased upon a ratio of the predicted output pulse amplitude error to anoutput pulse amplitude total value weighted by the input pulse width forthe PWM input signal. In another further embodiment, the predictiveerror correction signal can be based upon a ratio of the predictedoutput pulse amplitude error to an output pulse amplitude desired valueweighted by a pre-compensated pulse width for a prior PWM input signal.Still further, the amplitude error prediction circuitry can beconfigured to sense a varying or alternating current (AC) component of asupply voltage to predict the output pulse amplitude error for the PWMinput signal based on a prior PWM output signal. Alternatively, theamplitude error prediction circuitry can be configured to sense avarying or alternating current (AC) component of an output pulseamplitude for the PWM output signal to predict the output pulseamplitude error for the PWM input signal based on a prior PWM outputsignal.

As described below, other features and variations can be implemented andrelated methods and systems can be utilized, as well.

BRIEF DESCRIPTION OF THE DRAWINGS

It is noted that the appended drawings illustrate only exampleembodiments of the invention and are, therefore, not to be consideredlimiting of its scope, for the invention may admit to other equallyeffective embodiments.

FIG. 1A (prior art) is a block diagram for a prior solution that uses asupply voltage prediction signal to adjust the gain applied to PCM(pulse code modulated) digital input signals.

FIG. 1B (prior art) is a block diagram for a prior solution that uses afeedback signal based upon area comparison of PWM input/output signalsto adjust the edges of PWM signals.

FIGS. 2A, 2B, 2C and 2D are block diagrams for embodiments of switchingamplifiers including predictive feedback compensation (PFC) circuitryfor width adjustment.

FIG. 3 is a detailed block diagram for a switching amplifier includingpredictive feedback compensation (PFC) circuitry.

FIG. 4 is a signal diagram showing a deconstructed pulse with pulseerrors.

FIG. 5 is a circuit diagram for a negative edge delay cell that can beused to provide predictive feedback compensation.

FIG. 6A is a circuit diagram for a predictive feedback compensation(PFC) with open loop pulse width adjustment.

FIG. 6B is a timing diagram for the predictive feedback compensation(PFC) of FIG. 6A.

FIG. 7 is a circuit diagram for a linear interpolation predictor.

FIGS. 8A and 8B are circuit diagrams for a PWM switching amplifierincluding a predictive feedback compensation (PFC) having a linearinterpolation predictor and a feedback integrator.

FIGS. 9A and 9B are circuit diagrams for a differential embodiment for aPWM switching amplifier including a predictive feedback compensation(PFC) and feedback filter.

FIGS. 10A, 10B and 10C are block diagrams for embodiments of predictivefeedback compensations (PFCs) with closed loop pulse width adjustmentcircuitry.

FIG. 11 is an example timing diagram for the closed loop pulse widthadjustment circuitry of FIG. 10A.

FIG. 12 is a block diagram for a more general embodiment for the closedloop pulse width adjustment embodiments of FIGS. 10A, 10B and 10C.

DETAILED DESCRIPTION OF THE INVENTION

Methods and systems are disclosed for suppressing distortion and noisecaused by supply voltage variations and output amplitude switchingnon-idealities in PWM (pulse width modulated) switching amplifiers, suchas Class D digital audio amplifiers, through the use of predictivefeedback compensation.

As described herein, predictive feedback compensation (PFC) provides anapproach that uses output amplitude error information from the previouspulse frame(s) to predict how much to adjust the current pulse width tocorrectly compensate for gain non-idealities in the switching amplifiercaused by power supply voltage variations (and optionally other outputamplitude variations caused by switching non-idealities like variationsin r_(dson), which represents the resistance between the drain andsource of the output driver transistors). A performance benefit of thePFC approaches described herein is that the pre-compensating signal canbe used to correct non-idealities frame by frame, thereby helping toprevent the output signal from becoming corrupted in the first placewhile still not degrading the audio transient response of the open loopamplifier. Because power supply ripple corrupts the in-band output of aswitching amplifier non-linearly in a mixing, multiplicative fashion, itis beneficial to eliminate or attenuate the inter-modulation products inthe forward path. Overall performance may also be improved by adding afeedback loop to the PFC circuitry.

FIGS. 2A, 2B, 2C and 2D are block diagrams for embodiments of switchingamplifiers including predictive feedback compensation (PFC) circuitryfor adjusting the pulse width to mitigate PWM output pulse amplitudeerrors. In each of these embodiments, width adjustment circuitry 202receives an uncompensated PWM input signal and uses a predictive errorcorrection signal 206 to produce a PWM input signal to the output driverhaving a pre-compensated pulse width that has been adjusted forpredicted amplitude errors in the PWM output signals at the outputdriver. FIG. 2A uses the uncompensated PWM input signal (input to thewidth adjustment circuitry) and a predicted supply voltage error toprovide the predictive error correction signal 206. FIG. 2B uses thepre-compensated PWM signal (output of the width adjustment circuitry)and a predicted supply voltage error to provide the predictive errorcorrection signal 206. FIG. 2C uses the uncompensated PWM input signaland a predicted PWM output pulse amplitude to provide the predictiveerror correction signal 206. And FIG. 2D uses the pre-compensated PWMsignal and a predicted PWM output pulse amplitude to provide thepredictive error correction signal 206. Each of these embodiments is nowdiscussed in more detail.

FIG. 2A is a block diagram for a switching amplifier 200A includingpredictive feedback compensation (PFC) for pulse width adjustmentthrough the detection and prediction of amplitude errors using theuncompensated PWM input signal and a predicted supply voltage to providethe predictive error correction signal 206. As depicted, audio PCM inputsignals are received by a PWM controller 104, and the output of PWMcontroller 104 is provided to width adjustment circuitry 202. Widthadjustment circuitry 202 in turn provides width adjusted PWM signals todriver 106. Driver 106 then produces the PWM output signals, forexample, in the form of B-pulse (B) output signals and D-pulse (D)output signals for a Class D digital audio PWM switching amplifier.

The driver 106 is also coupled to receive power from supply voltage(Vp). The supply voltage (Vp), however, can have variations that lead toamplitude errors in the PWM output signals, and these errors translateinto distortion and noise in the audio output heard by a user for ClassD digital audio switching amplifiers. To compensate for these amplitudeerrors, amplitude error prediction circuitry 204 generates a predictiveerror correction signal 206 and provides it to the width adjustmentcircuitry 202.

The amplitude error prediction circuitry 204 receives the supply voltage(Vp) and outputs the predictive error correction signal 206. Theamplitude error prediction circuitry 204 also receives and utilizes thePWM input signals from PWM controller 104 for its error processing. Thewidth adjustment circuitry 202 and the amplitude error predictioncircuitry 204 form the predictive feedback compensation (PFC) 201. Ifdesired, optional feedback can also be provided. For example, a feedbacksignal from the PWM output signal (PWM_(OUT)) and a reference signalfrom the PWM input can be provided to feedback processing block 208. Thefeedback processing block 208 can compare the PWM input and PWM outputsignals and then provide a feedback error correction signal to the edgecorrection circuitry within the PFC 201. As such, the PFC approachdescribed herein can be used in conjunction with feedback systems.

FIG. 2B is a block diagram for a switching amplifier 200B includingpredictive feedback compensation for pulse width adjustment through thedetection and prediction of amplitude errors using the pre-compensatedPWM signal and a predicted supply voltage to provide the predictiveerror correction signal 206. In most respects, the embodiment 200B ofFIG. 2B is similar to the embodiment 200A of FIG. 2A. The differencebetween these two embodiments is that for embodiment 200B of FIG. 2B,the amplitude error prediction circuitry 204 receives thepre-compensated PWM signal that is provided as the output of the widthadjustment circuitry 202, rather than receiving the uncompensated PWMinput signal from the PWM controller 104, as is done for embodiment 200Ain FIG. 2A.

FIG. 2C is a block diagram for a switching amplifier 200C includingpredictive feedback compensation for pulse width adjustment through thedetection and prediction of amplitude errors using the uncompensated PWMinput signal and a predicted PWM output amplitude error to provide thepredictive error correction signal 206. In most respects, the embodiment200C of FIG. 2C is similar to the embodiment 200A of FIG. 2A. Thedifference between these two embodiments is that for embodiment 200C ofFIG. 2C, the amplitude error prediction circuitry 204 receives the pulseamplitude of the PWM output signal from the output of the drivercircuitry 106, rather than receiving the supply voltage (Vp), as is donefor embodiment 200A in FIG. 2A. It is further noted that amplitude errorprediction circuitry 204 can include sample-and-hold circuitry coupledto receive the pulse amplitude of the PWM output signal.

FIG. 2D is a block diagram for a switching amplifier 200D includingpredictive feedback compensation for pulse width adjustment through thedetection and prediction of amplitude errors using the pre-compensatedPWM signal and a predicted PWM output amplitude error to provide thepredictive error correction signal 206. In most respects, the embodiment200D of FIG. 2D is similar to the embodiment 200A of FIG. 2A. Onedifference between these two embodiments is that for embodiment 200D ofFIG. 2D, the amplitude error prediction circuitry 204 receives thepre-compensated PWM signal that is provided as the output of the widthadjustment circuitry 202, rather than receiving the uncompensated PWMinput signal from the PWM controller 104, as is done for embodiment 200Ain FIG. 2A. The other difference between these two embodiments is thatfor embodiment 200D of FIG. 2D, the amplitude error prediction circuitry204 receives the pulse amplitude of the PWM output signal from theoutput of the driver circuitry 106, rather than receiving the supplyvoltage (Vp), as is done for embodiment 200A in FIG. 2A. It is furthernoted that amplitude error prediction circuitry 204 can includesample-and-hold circuitry coupled to receive the pulse amplitude of thePWM output signal.

FIGS. 2A, 2B, 2C and 2D provide for compensation of the pulse widthbased upon output amplitude errors through the detection and measurementof voltage supply errors or direct measurement of the amplitude ofoutput pulse itself. As described further with respect to FIGS. 10A, 10Band 10C below, robustness of the width adjustment circuitry can beincreased using a closed loop system. For example, for embodiments 200Bin FIG. 2B and 200D in FIG. 2D, which use the width-adjusted,pre-compensated PWM signals output by the width adjustment circuitry 202to provide an input to the amplitude error prediction circuitry 204,stability can be realized by using a closed width adjustment loop, orsimilar circuit with servo feedback, examples of which are discussedwith respect to FIGS. 10A, 10B and 10C below. If the pre-compensated PWMsignals are used with an open loop adjustment circuit rather than aclosed loop adjustment circuit, the output width will tend to ratchet tothe maximum or minimum adjustment depending upon the sign of thepredicted error correction signal.

Embodiments for PWM switching amplifiers with PFC circuitry will now bediscussed below with respect to FIGS. 3, 4, 5, 6A, 6B, 7, 8A, 8B, 9A,9B, 10A, 10B and 10C.

PFC (Predictive Feedback Compensation) Approach

FIG. 3 is a detailed block diagram for a Class D audio switchingamplifier embodiment with the addition of PFC (predictive feedbackcompensation) blocks 201A and 201B to address output pulse amplitudeerrors caused by variations in the voltage supply as described herein.As depicted, PFC 201A and PFC 201B are positioned between PWM controller104 and switching amplifier 106. PFC 201A and PFC 201B achieve theadvantageous results described herein.

As depicted in this example, PCM digital audio input signals arereceived by a PCM-to-PWM converter 302, which outputs digital PWMsignals to a delta-sigma modulator 304. The output of delta-sigmamodulator 304 is a digital pulse width value (PWd_((N))) representingone PWM output frame that is sent to the PWM controller 104, which inturn produce PWM input signals for PFC 201A and PFC 201B, respectively.PFC 201A and PFC 201B produce the B and D PWM signals (PWMB, PWMD) thatare provided, respectively, to B-pulse timing control circuitry 312 andD-pulse timing control circuitry 314 within the switching amplifier 106.

The switching amplifier circuitry 106 takes the B/D PWM signals (PWMB,PWMD) and drives a desired load, such as a speaker 336. The B-pulsetiming control circuitry 312 produces output signals for gate drivers320 and 322. The gate drivers 320 and 322 provide control signals to thegates of PMOS drive transistor 342 and NMOS drive transistor 344,respectively, which in turn produce the B-pulse output signal applied tothe B-signal output pin (OUTB) 330. The D-pulse timing control circuitry314 produces output signals for gate drivers 324 and 326. The gatedrivers 324 and 326 provide control signals to the gates of PMOS drivetransistor 346 and NMOS drive transistor 348, respectively, which inturn produce the D-pulse output signal applied to the D-signal outputpin (OUTD) 332.

A passive LPF (low pass filter) 334 receives the B and D output signalsand provides output signals on nodes 352 and 354 to drive a speaker 336.The passive LPF 334 can include inductors and capacitors to providereconstruction filtering, such as inductors (L1) connected in the signalpaths between output pins 330 and 332 and output nodes 352 and 354,capacitors (C1) connected between the output nodes 352 and 354 andground, and a capacitor (C2) connected between the two output nodes 352and 354.

To produce a predictive error correction signal associated with thesupply voltage for the drive circuitry, PFC 201A is connected to receivethe supply voltage (Vp) for the output drive transistors 342 and 344. Asdepicted, PMOS drive transistor 342 has its source connected to thesupply voltage (Vp) and its drain connected to the output node thatconnects to pin 330. NMOS drive transistor 344 has its drain connectedto the output node that connects to pin 330 and its source connected toground (GND). PFC 201A is also configured to receive and utilize theB-pulse output signal (PWMB) from the PWM controller 104. PFC 201Aoperates to adjust the pulse width of the PWMB output signals to accountfor errors caused by variations in the supply voltage (Vp), as describedfurther below.

Similarly, to produce a predictive error correction signal associatedwith the supply voltage for the drive circuitry, PFC 201B is connectedto receive the supply voltage (Vp) for the output drive transistors 346and 348. As depicted, PMOS drive transistor 346 has its source connectedto the supply voltage (Vp) and its drain connected to the output nodethat connects to pin 332. NMOS drive transistor 348 has its drainconnected to the output node that connects to pin 332 and its sourceconnected to ground (GND). PFC 201B is also configured to receive anduse the D-pulse output signal (PWMD) from PWM controller 104. PFC 201Boperates to adjust the pulse width of the PWMD output signals to accountfor errors caused by variations in the supply voltage (Vp), as describedfurther below. It is further noted that the supply voltage (Vp) for theB-pulse output signal (PWMB) and the D-pulse output signal (PWMD) couldbe separate signals or be the same signal, as desired.

In operation of the open loop digital delta-sigma (ΔΣ) Class D switchingamplifier and passive LPF depicted in FIG. 3, each PCM digital inputsample is first converted to a digital PWM number representing thedesired output pulse width. The resulting high resolution multi-bitdigital PWM signal is then noise shaped and encoded by the PWMcontroller 104 into signal(s) for controlling the output state of theswitching amplifier. For single-ended configurations, this will be asingle PWM signal whereas for BTL (bridge-tied load) configurations asshown in FIG. 3, this can be a pair of PWM signals (PWMB and PWMD), onefor each side of the bridge. The width of the switched output pulse ateach output pin 330 and 332 is determined by the width of the input PWMcontrol signal, and the amplitude of the switched output pulse isdetermined by the switching amplifier supply voltage level (Vp).Variations in the supply voltage (Vp) and in edge transitions introduceerrors in the filtered output signals which generally correlate to acontinuous integration of the area under the output pulse signals. It isthe amplitude errors caused by variations in the supply voltage (Vp)that PFC 201A and PFC 201B address.

It is noted that PFC 201A and PFC 201B receive the supply voltage (Vp)in embodiment 300 of FIG. 3. As with the embodiments in FIGS. 2C and 2D,the PFC 201A and the PFC 201B could instead receive the pulse amplitudeof the PWM output signals applied to pins 330 and 332. In addition, thePFC 201A and the PFC 201B could receive both the supply voltage (V_(p))and the pulse amplitude of the PWM output signals, if desired.

FIG. 4 is a signal diagram showing pulse area errors. In particular,FIG. 4 shows a deconstruction of these errors for a single pulse frameinto a time-based error component 406 and a voltage-based (oramplitude-based) error component 404, for a desired or reference pulsearea 402. As noted above, the amplitude of the output pulse willcorrelate to the voltage supply received by the output driver. Lookingback to FIG. 4, the amplitude for the desired or reference pulse isrepresented by the term V_(r), and the pulse width for the referencepulse is represented by the term PW_(i). The output amplitude for thepulse is represented by the term V_(o), and the output width isrepresented by the term PW_(o). Using these designations, the totalerror in the output pulse area can represented by an amplitude(voltage-based) error (E_(V)=(V_(o)−V_(r))*PW_(i)) plus a width(time-based) error (E_(W)=(PW_(o)−PW_(i))*V_(o)), according to thefollowing equation:

E _(TOTAL) =E _(V) +E _(W)=[(V _(o) −V _(r))*PW_(i)]+(PW_(o)−PW_(i))*V_(o) =V _(o)*PW_(o) −V _(r)*PW_(i)

The goal of the predictive feedback compensation described herein is togenerate an output pulse having the same area as the desired orreference pulse area (V_(r)*PW_(i)) by correcting for the amplitude(voltage-based) error represented by area 404.

If the total error (E_(TOTAL)) is set to zero, thenV_(o)*PW_(o)=V_(r)*PW_(i). The precompensated pulse width (PW_(o)) isthe ideal pulse width (PW_(i)) plus the pulse width correction (PW_(c)),resulting in the representation:

V _(o)*(PW_(i)+PW_(c))=V _(r)*PW_(i).

Solving for PW_(c) in terms of PW_(i) (per examples of FIGS. 2A and 2C)results in the following expression:

PW_(c)=(PW_(i) *V _(r) /V _(o))−PW_(i)=PWi*[(V _(r) −V _(o))/V_(o)].

It is noted that the desired or reference pulse amplitude V_(r) in mostpractical applications will typically be the DC component of the outputamplitude V_(o), and the ripple (or AC) component (V_(n)) will typicallybe the difference between the output amplitude absolute voltage (V_(o))and this desired or reference pulse amplitude (V_(r)) (i.e.,V_(n)=V_(o)−V_(r)). The output amplitude absolute voltage (V_(o)) canalso be estimated using the supply absolute voltage (V_(p)) (i.e.,V_(p)=V_(o)=V_(n)+V_(r)). Substituting and rearranging terms results ina feedforward algorithm:

PW_(c)=−PW_(i) *V _(n)/(V _(n) +V _(r))

PW_(c)=−PW_(i) *V _(n) /V _(p).

Thus, the effects of power supply ripple can be eliminated bypre-compensating the input pulse width (PW_(i)) with a counteractingadjustment (PW_(c)) proportional to input pulse width (PW_(i)) times theratio of ripple voltage (V_(n)) to supply voltage (V_(p)=V_(n)+V_(r)).For this solution that follows the examples of FIGS. 2B and 2D, theripple voltage (V_(n)) represents the predicted output pulse amplitudeerror for the PWM input signal based upon a prior PWM output signal. Andthe supply voltage (V_(p)=V_(n)+V_(r)) represents an output pulseamplitude in the form of an output pulse amplitude total value. In thisway, the pre-compensating operation pre-compensates the input pulsewidth for the PWM input signal with an adjustment based upon a ratio ofthe predicted output pulse amplitude error (V_(n)) to the predictedoutput pulse amplitude total value (V_(p)) weighted by the input pulsewidth (PW_(i)) for the PWM input signal to produce the pre-compensatedpulse width for the PWM input signal.

Alternatively, the counteracting adjustment (PW_(c)) can be solved interms of the corrected pulse width output PW_(o), whereinPW_(o)=PW_(i)+PW_(c) (per examples of FIGS. 2B and 2D). The resultingequation is:

V _(o)*PW_(o) =V _(r)*(PW_(o)−PW_(c)).

Again, substituting and rearranging results in a feedback algorithm:

PW_(c)=PW_(o)−(PW_(o) *V _(o) /V _(r)), or

PW_(c)=PW_(o)*[1−(V _(o) /V _(r))], or

PW_(c)=PW_(o)*[(V _(r) −V _(o))/V _(r)], or

PW_(c)=−PW_(o)*(V _(n))/(V _(r)).

Thus, alternatively, the effects of power supply ripple can beeliminated by pre-compensating the input pulse width (PW_(i)) with acounteracting adjustment (PW_(c)) proportional to corrected pulse width(PW_(o)) times the ratio of ripple voltage (V_(n)) to the reference (ordesired) voltage (V_(r)). For this alternative solution that follows theexamples of FIGS. 2A and 2C, the ripple voltage (V_(n)) represents thepredicted output pulse amplitude error for the PWM input signal basedupon a prior PWM output signal. And the reference or desired voltage(V_(r)) represents an output pulse amplitude in the form of an outputpulse amplitude desired value. In this way, the pre-compensatingoperation pre-compensates the input pulse width (PW_(i)) for the PWMinput signal with an adjustment based upon a ratio of the predictedoutput pulse amplitude error (V_(n)) to the output pulse amplitudedesired value (V_(r)) weighted by a pre-compensated pulse width (PW_(o))for a prior PWM input signal to produce the width adjustment (PW_(c))for the PWM input signal.

two alternative pre-compensating techniques set forth above can berepresented more generally using EQUATION 1 below. Using EQUATION 1,pre-compensating operation pre-compensates the input pulse width(PW_(i)) for the PWM input signal with a width adjustment (PW_(c)) basedupon a ratio of the predicted output pulse amplitude error (V_(n)) to anoutput pulse amplitude (V=V_(p) or V_(r)) weighted by a pulse width(PW=PW_(i) or PW_(o)). In other words, the equation related to FIGS. 2Aand 2C, which is:

PW_(c)=−PW_(i) *V _(n) /V _(p)   [EQUATION 2A]

and the equation related to FIGS. 2B and 2D, which is:

PW_(c)=−PW_(o) *V _(n) /V _(r)   [EQUATION 2B]

can be expressed more generally as the following:

PW_(c)=−PW*V _(n) /V   [EQUATION 1]

where PW is a pulse width for a PWM signal and V is a related outputpulse amplitude. For EQUATION 2A above, PW is the input pulse width(PW_(i)) for the prior PWM output signal, and V is the related outputpulse amplitude total value (V_(p)). And for EQUATION 2B above, PW isthe pre-compensated pulse width (PW_(o)) for a prior PWM input signal,and V is the related output pulse amplitude desired value (V_(r)).

It is further noted that the predicted output pulse amplitude error(V_(n)) for EQUATION 1, EQUATION 2A and EQUATION 2B can be predictedusing the supply voltage driving the output switching amplifiers and/orusing the amplitude of the output PWM signal itself. FIGS. 2A and 2Bprovide example embodiments where the supply voltage is used for theamplitude error prediction circuitry. As such, measuring a varying oralternating current (AC) component of the supply voltage is used topredict the output pulse amplitude error (V_(n)) for the PWM inputsignal based on a prior PWM output signal. FIGS. 2C and 2D provideexample embodiments where the output PWM signal is used for theamplitude error prediction circuitry. As such, measuring a varying oralternating current (AC) component of the output pulse amplitude for thePWM output signal is used to predict the output pulse amplitude error(V_(n)) for the PWM input signal based on a prior PWM output signal.

Compensating Circuit Structure—Negative Edge Delay Cell

FIG. 5 is a circuit diagram for a negative edge delay cell that can beused to implement width adjustment circuitry for predictive feedbackcompensation. This circuit structure 500 implements delay for a fallingedge with the voltage ratio relationship set forth in the equationsabove; however, as described herein either one or both edges of thepulse could be adjusted, as desired. As depicted, an input PWM signal(PWM_(i)) 502 is applied to the gate of MOS transistor 506, which hasits source connected to ground and its drain connected to node 508. Thecapacitor (C_(t)) is coupled between node 508 and ground. Node 508 isalso connected to a voltage-to-current (G_(m)) block 512. The supplyvoltage (V_(p)), which represents the estimated amplitude of the outputsignal (V_(o)) in the equations above, is provided as the voltage inputto the voltage-to-current (G_(m)) block 512 and then as a current tonode 508 to charge capacitor (C_(t)). The voltage on node 508 is thenconnected to an input of comparator 510, which also receives a thresholdvoltage (V_(t)) as an input. The threshold voltage (V_(t)) can be madeto be proportional to the supply absolute voltage (V_(p)) plus a biasvoltage (V_(b)). The comparator 510 will output a high level if node 508is below the threshold voltage (V_(t)) and a low level if node 508 isabove the threshold voltage (V_(t)). The comparator 510 operates toproduce an output PWM signal (PWM_(d)) 504.

It is noted that the supply absolute voltage (V_(p)) is used here torepresent the estimated amplitude of the output signal (V_(o)). And thedesired or reference voltage (V_(r)) is used to represent the desired orreference amplitude of the output signal.

For the negative edge delay cell 500 of FIG. 5, the delay (τ_(df)) 520between the input falling edge of the input PWM signal (PWM_(i)) 502 andoutput falling edge of the output PWM signal (PWM_(d)) 504 is given bythe following equation:

τ_(df)=(C _(t) /G _(m))*V _(t) /V _(p)+τ_(a),

-   -   where τ_(a) represents the delay of the comparator propagation.        If the threshold voltage (V_(t)) is made inversely proportional        to the power supply ripple (or AC) voltage (V_(n)) weighted by        factor (α) proportional to the desired pulse width is added to a        bias voltage (V_(b)), then the following expression can be made:

V _(t) =−α*V _(n) +V _(b)=−α*(V _(p) −V _(r))+V _(b),

the delay can be seen to be:

τ_(df)=−(C _(t) /G _(m))*α*V _(n) /V _(p)+(C _(t) /G _(m))*V _(b) /V_(p)+τ_(a),

Which is of the form given above for PW_(c) plus a bias latency, where

PW_(i)=(C _(t) /G _(m))*α

is a weighting factor for scaling the correction proportional to thedesired pulse width, and the bias latency (τ₁) is given by:

τ₁=[(C _(t) /G _(m))*V _(b) /V _(p)]+τ_(a)

If the bias voltage (V_(b)) is set proportional (or equal) to thevoltage controlling the capacitor ramp current (e.g., V_(p) throughblock 512), the bias latency to a first order will be time invariant anddetermined by the RC time constant plus the comparator propagation delay(τ_(a)).

With this circuit structure of FIG. 5, a sub-system can be configuredfor robustly compensating the PWM signal for variations in supplyvoltage or output pulse amplitude. One possible implementation toachieve this result is depicted with respect to FIG. 6A and the timingdiagram of FIG. 6B.

Single-Ended Circuit Solution Embodiment

FIG. 6A is a circuit diagram for a single-ended embodiment for apredictive feedback compensation embodiment 600 with open loop-pulsewidth adjustment and using the negative edge delay cell of FIG. 5. FIG.6B is a timing diagram for the predictive feedback compensation of FIG.6A. As can be seen from inspection of the sub-system diagram in FIG. 6A,the upper negative edge delay cell is configured to delay the risingedge and the lower cell is configured to delay the falling edge. Theoutput transitions for S-R latch 604, as shown in FIG. 6B, are inresponse to the delayed edge in each case, resulting in an output pulse(PWMpc) with a rising edge delayed by τ₁+τ_(dr) and a falling edgedelayed by τ₁+τ_(df), where τ₁ is the bias latency, where τ_(dr) is therising edge delay and where τ_(df) is the falling edge delay.

As depicted in the embodiment 600 of FIG. 6A, the uncompensated PWMinput signal (PWM_(i(T))) 502 is received and sent to the S-input of theS-R latch 602A. The inverted output (QB) of the S-R latch 602A isprovided to the gate of transistor 506A, which is part of the uppernegative edge delay cell. The node 508A is provided to the comparator510A. And output 520A from comparator 510A is then provided through aninverter as signal 622 to the S-input of output S-R latch 604. Thenon-inverted output (Q) of the S-R latch 604 is the pre-compensated PWMinput signal (PWMpc) 620 that has had the time (T) of its pulse widthadjust by a correction factor (ΔT) so that the new pulse width is T-ΔT,as discussed in more detail below. The difference between theuncompensated PWM input signal (PWM_(i(T))) and the pre-compensated PWMinput signal (PWMpc) 620 would represent the pulse width adjustment usedto compensate for amplitude errors.

As depicted in the embodiment 600 of FIG. 6A, the uncompensated PWMinput signal (PWM_(i(T))) 502 is also sent through an inverter as signal601 to the S-input of the S-R latch 602B. The inverted output (QB) ofthe S-R latch 602B is provided to the gate of transistor 506B, which ispart of the lower negative edge delay cell. The node 508B is provided tothe comparator 510B. And output 520B from comparator 510B is thenprovided through an inverter as signal 623 to the R-input of output S-Rlatch 604.

It is noted that the non-inverted output (Q) and the inverted output(QB) of the S-R latch 604 is also sent back to be the R-inputs of S-Rlatch 602A and S-R latch 602B, respectively. It is further noted thatthe two input S-R latches 602A and 602B enable operation with narrowpulses by preventing the trailing edge from discharging the rampingcapacitors (C_(t)) before the delayed output transitions. Theseadditional S-R latches are not required for fundamental operation, butdoes allow operation to the maximum modulation index while helping toprevent inadvertent pulse swallowing.

As discussed with respect to FIG. 5, the supply absolute voltage (V_(p))is provided to nodes 508A and 508B through voltage-to-current (G_(m))blocks 512A and 512B in each delay cell. It is further noted that asample-and-hold (S/H) block 614 has also been included beforevoltage-to-current (G_(m)) blocks 512A and 512B to capture the supplyabsolute voltage (V_(p)) at desired points of time during the operationof the circuitry.

The threshold voltages (V_(t)) for comparator 510A and comparator 510Bare generated from the supply absolute voltage (V_(p)) 618 usingcircuitry 630. Circuitry 630 acts as the amplitude error predictioncircuitry in this embodiment. Supply absolute voltage (V_(p)) 618includes both a DC (desired or reference voltage—V_(r)) component and anAC (ripple voltage—V_(n)) component as discussed above. In oneembodiment the, supply absolute voltage (V_(p)) 618 is provided to ahigh pass filter (HPF) 606 that filters out the DC component. Forexample, a HPF 606 that rejects frequencies below about 20 Hz can beused to pass the ripple or AC component (V_(n)) of the supply absolutevoltage (V_(p)). The output (V_(n)) of HPF 606 is then provided to block608 that is configured to provide a weighted-integrate-and-dump functionon the ripple (or AC) component (V_(n)) of the supply voltage (V_(p))using the pulse width timing of the uncompensated PWM input signal(PWM_(i(T))) 502. The output of block 608 is provided through asample-and-hold (S/H) block 616 as a positive input to summation block610A and as a negative input to summation block 610B. Supply absolutevoltage (V_(p)) is also provided through sample-and-hold (S/H) block 614as positive inputs to summation blocks 610A and 610B to set the biaslatency for the PFC. The output of summation block 610A is provided asthe threshold voltage (V_(t)) input to comparator 510A, and the outputof summation block 610B is provided as the threshold voltage (V_(t))input to comparator 510B.

FIG. 6B is a timing diagram 650 for the embodiment 600 of FIG. 6A.Represented in timing diagram 650 is the input PWM signal (PWM_(i(T)))502, inverted input PWM signal (PWMi_((T)) _(—) bar) 601, the outputsignal (S_(out)) 622 provided to the S-input of S-R latch 604, theoutput signal (R_(out)) 623 provided to the R-input of S-R latch 604,and the pre-compensated PWM input signal (PWMpc) 620 from the Q output(Q_(out)) from S-R latch 604. As depicted, dotted line 652 representsthe rising edge timing (t_(i)) for the original input PWM pulse. Dottedline 654 represents the position of the rising edge if moved solely dueto the fixed timing latency (t_(latency)) of the PFC circuitry 600.Dotted line 656 represents the falling edge timing (t_(f)) for theoriginal input PWM pulse. And dotted line 658 represents the position ofthe falling edge if moved solely due to the fixed timing latency(t_(latency)) of the PFC circuitry 600. As shown, the pre-compensatedPWM input signal (PWMpc) 620 has had its rising edge delayed by +ΔT/2from the latency only timing and has had its falling edge sped up byΔT/2 from the latency only timing so that the entire pulse width (T) hasbeen narrowed by a total of ΔT to produce an output width of T-ΔT, asindicated above and discussed in more detail below.

It is noted that the delay latency (τ_(latency)) includes the comparatorpropagation delay and a constant delay set by bias threshold (V_(t)) onthe delay cell comparators 510A and 510B. By setting this bias thresholdvoltage (V_(t)) proportional (or equal) to the power supply absolutevoltage (V_(p)) that is also setting the current in the timingcapacitors (C_(t)), variations in the voltage track out and the constantdelay portion of the delay latency depends only on the (C_(t)/G_(m))time constant. Preferably, the absolute value of the latency is madelarge enough to provide compensation for the maximum peak to peakvariation in the supply voltage (V_(p)).

It is further noted that while FIG. 6A shows two voltage-to-currentconverters, this is merely illustrative convenience for clarifying thecircuit operation. Because the capacitor charging currents areconfigured to be identical for the upper and lower path, a singlevoltage-to-current converter can be used with a dedicated current mirrorleg for each charging capacitor.

Differential Mode Operation

In a further embodiment, the rising and falling edge delay differencesare configured to move in equal and opposite directions proportional tothe ripple (or AC) component of the supply absolute voltage (V_(p))weighted by the pulse width. Employing this type of differential edgedelay scheme (e.g., as opposed to a single edge scheme) helps maintainthe relative position of pulse centers between the input PWM pulse andthe adjusted pre-compensated pulse. In alternative embodiments, thegeneral scheme can also be employed whereby only one edge moves whilethe other remains relatively fixed. Still further, both edges can bemoved, but by different amounts.

One way to generate the ripple (or AC) component (V_(n)) of the supplyabsolute voltage (V_(p)) is by using a high pass filter, such as HPF 606in FIG. 6A. Another approach to obtain this ripple (or AC) component ofthe supply absolute voltage (V_(p)) is to subtract from the supplyabsolute voltage (V_(p)) a fixed reference voltage representing the DCcomponent of the supply absolute voltage (V_(p)) For example, a fixedreference could be the supply absolute voltage (V_(p)) filtered by a lowpass filter, or it could be a locally generated voltage. Any staticoffset between this reference voltage and the actual average outputstage supply voltage, however, will result in a static gain adjustmentin the output stage degraded power supply rejection performance as setforth below.

One convenient way to weight the supply ripple (or AC) voltageproportional to the pulse width is with a simple integrator such thatthe output amplitude error prediction is given by:

ΔV _(t) =α*V _(n)=[τ_(i)/(C _(i) *R _(i))]*V _(n)

-   -   where τ_(i)=input pulse width per frame, and

where Ci and Ri are the integration capacitor and resistor,respectively.

This presents a small real time problem since the leading edge must bedelayed before the current pulse width is known. Therefore, it isnecessary to estimate, or predict, the current pulse width and a ripple(or AC) component of the power supply voltage based on previous values.For simplicity, it can be assumed that the pulse width and supply ripplevoltage component associated with the previous pulse is a good predictorfor the current values. In FIG. 6A, this simple predictor is implementedfor the ripple (or AC) signal and the supply absolute voltage V_(p) witha sample-and-hold circuit triggered off the falling edge of the PWMipulse. While this assumption results in good performance, the predictioncan be significantly improved with the enhancement discussed in moredetail below.

Pulse Width Correction Analysis

Ultimately, the comparator threshold voltage (V_(t)) includes twocomponents: the first being a prediction of the supply absolute voltageV_(p) and the second being a prediction of the power supply ripple (orAC) voltage V_(n) weighted by the pulse width using an integrator block.The sum of these two components is applied to the V_(t) of the risingedge delay cell by block 610A, and the difference of these twocomponents is applied to the V_(t) of the falling edge cell by block610B. The first component sets a common mode delay for the rising andfalling edges, and the second component sets a differential mode delaythat symmetrically modulates the pulse width, where the rising edgedelay (τ_(dr)) and the falling edge delay (τ_(df)) can be represented bythe following:

τ_(dr)=[(C _(t) /G _(m))/(C _(i) *R _(i))]*τ_(i)*(V′ _(n) /V′ _(p))+C_(t) /G _(m)+τ_(a),

τ_(df)=−[(C _(t) /G _(m))/(C _(i) *R)]*τ_(i)*(V′ _(n) /V′ _(p))+C _(t)/G _(m)+τ_(a),

where V′_(n) and V′_(p) represents the estimated values of the voltagesV_(n) and V_(p). Thus, the adjustment in the output pulse width isdefined as the difference in delay between the falling and rising edges,

Δτ_(d)=τ_(df)−τ_(dr)=−[(2*C _(t) /G _(m))/(C _(i) *R _(i))]*τ_(i)*(V′_(n) /V′ _(p))

Δτ_(d)=τ_(df)−τ_(dr) =−K*τ _(i)*(V′ _(n) /V′ _(p)),

where K=2*(C _(t) /C _(i))*(1/(G _(m) *R _(i)))

and PW_(i) =K*τ _(i)

This is exactly the form required for perfectly cancelling the effect ofpower supply variations at the output of the switching amplifier, whilemaintaining the relative pulse center position. The resulting filteredoutput signal Vo can be described by the following relationship, where Tis the PWM frame period:

Vo=[(τ_(i)+Δτ_(d))/T]*V _(p)

Vo=(τ_(i) /T)*V _(p)*(1+Δτ_(d) /τ _(i))

Vo=(τ_(i) /T)*V _(p)*(1−K*V′ _(n) /V′ _(p))

Vo=(τ_(i) /T)*(V _(p) −K*V _(n) *V _(p) /V′ _(p))

Vo=(τ_(i) /T)*(V _(r) +V _(n) −K*V′ _(n) *V _(p) /V′ _(p))

Vo=(τ_(i) /T)*{V _(r) +V _(n)*[1−K*(V′ _(n/n))*(V _(p) /V′ _(p))]},

which shows that the power supply amplitude variation is attenuated bythe factor

α=1−K*(V′ _(n) /V _(n))*(V _(p) /V′ _(p)),

-   -   Where K is the product of three ratios given by

K=2*(C _(t) /C _(i))*(1/(G _(m) *R _(i))).

For ideal component matching and prediction, α=0 and perfectcancellation results.

Component Mismatch and Prediction Errors

Given “r” is the component matching tolerance, error attributable tocomponent mismatch is on the order of (1-r)³, assuming pessimisticallythat the mismatches between resistors, capacitors and current mirrorsare correlated. If r=0.1% for example, the power supply variation wouldbe attenuated by more than 50 dB.

It is reasonable to assume that prediction error results primarily fromestimating the supply ripple because the ripple will generally be afraction of the supply absolute voltage. Therefore for a given absoluteprediction error for the voltage level, the percentage mismatch for(V_(p)/V′_(p)) will be a fraction of the percentage mismatch for(V′_(n)/V_(n)).

Using only the previous sample of the weighted V′_(n) to predict thenext sample results in a prediction error equal to how much the powersupply might change from sample to sample. For a tone, the maximum erroroccurs when the signal is at zero since this is the point of maximumrate of change. For a tone of frequency f_(m) and PWM frame rate off_(c), the error will be sin(2*π*f_(m)/f_(c)). Assuming f_(c)=920 kHz,the resulting power supply ripple attenuation with full scale PWMmodulation will be −43 dB for a 1 kHz tone and −29 dB for a 5 kHz tone.

This performance can be greatly improved upon by linearly interpolatingthe previous two samples to predict the next sample. This operation canbe implemented with a 2× gain block and a sample/hold circuit.

FIG. 7 provides a circuit diagram for an embodiment 700 for such alinear interpolation predictor. An input signal 702 is provided tosample-and-hold (S/H) circuitry 706 and to 2× gain block 704. The outputof the gain block 704 is provided as a positive input to summation block708, and the output of sample-and-hold (S/H) circuitry 706 is providedas negative input to summation block 708. The output of summation block708 provides the output signal 710 for the linear interpolationpredictor. The linear interpolation predictor 700 can be inserted inplace of the sample-and-hold (S/H) circuitry 616 in FIG. 6A.

When the linear interpolation predictor 700 of FIG. 7 is applied inplace of the sample-and-hold (S/H) circuitry 616 in FIG. 6A, the outputamplitude error prediction value ΔV_(t) produced by this circuitry canbe expressed as:

ΔV _(t)=τ_(i) *V _(ni)=τ_(i)*(2*V _(n(i−1)) −V _(n(i−2))),

-   -   where V_(nk) represents the output pulse amplitude estimation at        sample time k.

The maximum error will occur when the rate of change of the signalderivative is maximum, which occurs at the signal peak, and can be shownto be 2*(1−cos(2*π*f_(m)/f_(c))). Again assuming f_(c)=920 kHz, theresulting power supply ripple attenuation with full scale PWM modulationwill be −86 dB for a 1 kHz tone and −58 dB for a 5 kHz tone.

In the case of a double sided BD modulated PWM signal (described in moredetail below) in which the differential pulse width is what iscompensated, the effective f_(c) is double and the resulting attenuationis −98 dB for a 1 kHz tone and −70 dB for a 5 kHz tone.

Using the two previous samples to estimate the next provides sufficientprediction accuracy that component matching becomes the dominantlimiting factor for PSR enhancement using this technique.

It is further noted that static DC voltage offset mismatch betweenthreshold voltages (V_(t)) or the comparators 510A and 510B do notimpact the PSR (power supply rejection) attenuation, but do introduce aDC offset (V_(os)) at the output expressed by the following:

V _(os)=τ_(os) *V _(p) /T

V _(os) =V _(tos) *C _(t)/(T*G _(m)),

-   -   Where τ_(os) is the delay offset caused by a comparator        threshold voltage offset of V_(tos).

Assuming a common mode latency C_(t)/G_(m)<10% of the PWM frame rate(this is a reasonably nominal assumption although dependent on specificsystem requirements and design choices), the resulting open loop outputDC offset will be 5% of the delta offset between the comparators.

Jitter and Noise Considerations

It is noted that the primary design concerns for the blocks in FIG. 6Aare component matching and minimizing noise induced jitter. Componentmismatch in and of itself only affects the attainable power supplyripple attenuation and does not degrade the desired PWM signalintegrity. On the other hand, noise induced jitter will not affect theattainable attenuation but will degrade desired PWM signal SNR(signal-to-noise ratio). As with any circuits through which thecritically timed PWM signals pass, care must be taken to minimize noiseinducing jitter on the transition edges. While most of this is circuitdesign, the one system design consideration is to minimize the commonmode latency so that the charging capacitor ramp is as steep aspossible. While it should not be necessary in general, one could chooseto trim the voltage-to-current converter resistor R_(m) atpost-manufacturing test to achieve greater control over the common modelatency and the charging, capacitor ramp time.

Combining PFC Circuitry with Traditional Feedback Circuitry

Using PFC circuitry as described herein, the power supply rejection ofthe open loop forward path in a Class D switching amplifier can beimproved by more than 50 dB. Even so, other time and amplitude basednon-idealities and non-linearities are potentially left unchecked.Therefore, it is desirous to add classical feedback in addition to PFCto correct for these residual errors and further enhance the performanceof the amplifier.

In fact, the same pulse edge delay cell used for PFC can also be usedfor feedback control by summing the integrated feedback control errorsignal with the output of the predictive integrator. This will cause thepulse width to incrementally adjust in an effort to drive theinstantaneous error signal to zero.

FIGS. 8A and 8B together provide a circuit diagram for an embodiment 800of a predictive feedback compensation including a linear interpolationpredictor and a feedback integrator. The predictive feedbackcompensation 802 is configured the same as the embodiment 600 of FIG.6A, except that linear interpolation predictor 700 has been inserted inplace of the sample-and-hold (S/H) circuitry 616. In addition, asingle-ended switching amplifier 804 is also depicted that receives thepre-compensated PWM input signal (PWMpc) from the predictive feedbackcompensation 802. The switching amplifier 804 is configured the same asthe B-pulse portion of switching amplifier 106 in FIG. 3. And similar tothe embodiment of FIG. 3, the switching amplifier 804 drives a speakerwith an output voltage (V₀) generated by passing an output PWM signal(PWM_(o)) having a pulse width (T) corrected by ΔT throughreconstruction low pass filter circuitry including inductor L1 andcapacitor C1.

Also depicted in FIGS. 8A and 8B is a feedback filter 806 that providesa feedback error signal (V_(e)) to the predictive feedback compensation802. In particular, as depicted, the feedback integrator 806 includes adifference amplifier 808, a loop filter (−H(s)) 810, and sample-and-hold(S/H) circuitry 812. The difference amplifier 808 has its positive inputcoupled to ground and feedback capacitor C_(F) coupled between itsoutput and its negative input. The negative input is further coupled tothe inverted input PWM signal (PWM_(i) _(—) bar) through resistor R_(F2)and to the output PWM signal (PWM_(o)) through resistor R_(F1). Thesecombined connections act to create a difference signal between the PWMinput signal (PWM_(i)) and PWM output signal (PWM_(o)) at the negativeinput of amplifier 808. The output of difference amplifier 808 passesthrough loop filter (−H(s)) 810 and then through sample-and-hold (S/H)circuitry 812 to produce the error feedback signal (V_(e)) 814. Thiserror feedback signal (V_(e)) 814 is then coupled to PFC 802 as anadditional positive input to the summation block 708 within the linearinterpolation predictor 700. It is noted, however, that embodiment 800illustrates one way to integrate classical feedback with the PFC 802.Other feedback techniques could also be used.

In operation of the feedback filter 806, the integral of the differencebetween a level attenuated output pulse sequence (PWM_(o)) and the inputreference pulse sequence (PWM_(i)) formed by amplifier 808 to create aninstantaneous error signal. This instantaneous error signal issubsequently filtered by a loop filter (−H(s)) 810 to create thefeedback control error signal. This feedback control error signal canalso be pass through S/H circuitry 812 before being provided as theerror feedback signal (V_(e)) to the PFC 802. If the error signal Ve ispositive, meaning the area of the output pulse sequence (PWM_(o)) islarger than the area of the input pulse sequence (PWM_(i)), the pulseedge delay cell will decrease the pulse width until the area of theoutput pulses equal that of the input pulses and the instantaneous errorsignal is zero. It is noted that the error signal Ve could also benegative, meaning the area of the output pulse sequence (PWM_(o)) issmaller than the area of the input pulse sequence (PWM_(i)).

Advantageously, the feedback superimposes nicely onto the predictivefeedback compensation without any mutual interference. The PFCeliminates most output error caused by power supply ripple, reducing thetotal amount of correction required by the traditional feedback network.

Application To Differential BTL Outputs

The preceding discussion dealt primarily with single-ended switchingamplifier embodiments. The following discussion is directed todifferential embodiments, an example for which is provided in FIGS. 9Aand 9B.

In general, for differential BTL (bridge-tied load) applications, thesingle-ended implementation can be used for each signal, and the pulsewidth compensation will translate to the differential signal. However,this technique can also introduce an unwanted differential pulse phasemodulation (i.e., pulse position shifting). In addition, if the commonmode phase varies with respect to the differential mode (as in the caseof common mode carrier suppression), the differential pulse widths willbe modulated by the common mode phase variation. This is likely alsotrue for any pulse width compensating scheme.

Therefore, for those applications where it is desired not to affect thedifferential mode phase, it may be preferred to implement the feedbackand the PFC exclusively in differential space in such a way that thedifferential pulse center positions remain invariant. This requires thatthe respective pulse edges that define the differential pulse compensatein equal but opposite directions. In most cases for BD modulation, thesedefining pulse edges are associated with two distinct single-ended PWMsignals and therefore the compensation must be coordinated between thetwo PWM signal edges.

Because of the symmetry of the BTL output for positive and negativesignals, the required direction of compensation will be oppositedepending on the sign of the signal. This complication is minor and asan example can be accommodated by using signal sign information toreverse the compensation direction for the edges of each respective PWMpulse. The sign information can be provided by the modulator orlogically divined by comparing the positive/negative (P/N) orpulse-B/pulse-D (B/D) PWM signals.

FIGS. 9A and 9B together provide a circuit diagram for a differentialembodiment 900 for a predictive feedback compensation (PFC) 902including a feedback integrator 906. As depicted, the PFC 902 includes afirst circuit configured similar to PFC embodiment 802 of FIG. 8 thatreceives a positive differential input PWM signal (PWM_(bi)(T)) 901 andproduces a first pulse width adjusted output signal PWM_(bpc). PFC 902also includes a second circuit configured similar to PFC embodiment 802of FIG. 8 that receives a negative differential input PWM signal(PWM_(di)(T)) 903 and produces a second pulse width adjusted signalPWM_(dpc). It is noted that the positive PWM signal (PWM_(bpc)) 930 andthe negative PWM signal (PWM_(dpc)) 932 correlate to the B-pulse PWMsignal (PWMB) and D-pulse PWM signal (PWMD) discussed in FIG. 3.

Further additions to the PFC 902 for the differential embodiment 900include the XOR block 960, the sign (SGN) input signal 922, and the MUXs954 and 956. The XOR block 960 receives the positive differential inputPWM signal (PWM_(bi)(T)) 901 and the negative differential input PWMsignal (PWM_(di)(T)) 903 and then provides an XORed output signal to theweighted-integrate-and-dump circuitry. The sign (SGN) signal 922controls the MUXs 954 and 956 which receive outputs from each of thesummation circuits that provide the threshold voltages (V_(t)) to thecomparators in PFC 902.

The B-pulse PWM signal (PWM_(bpc)) 930 and the D-pulse PWM signal(PWM_(dpc)) 932 are provided to switching amplifiers 904A and 904B. Thisswitching amplifier circuitry has been discussed above with respect toswitching amplifier 106 in FIG. 3 and switching amplifier 804 in FIG. 8.As with the embodiment 300 of FIG. 3, the outputs from the switchingamplifiers 904A and 904B are sent through reconstruction low pass filter(LPF) circuitry (L1, C1, C2) to drive an output device (e.g. speaker336). As shown in FIG. 9, the output signal (PWMB_(o(T−ΔT))) ofswitching amplifier 904A has a pulse width of the base width (T) minusthe pulse width adjustment (ΔT). And the output signal (PWMD_(o(T+ΔT)))of switching amplifier 904B has a pulse width of the base width (T) plusthe pulse width adjustment (ΔT).

The feedback filter 906 is similar to feedback filter 806 of FIG. 8. Inthis embodiment, however, the difference amplifier 907 receivesadditional signals at its inputs. At its positive input, differenceamplifier 907 receives output signal (PWMB_(o(T−ΔT))) from switchingamplifier 904A through a resistor and the input signal (PWM_(bi(T))) 901through a resistor. At its negative input, difference amplifier 907receives the output signal (PWMD_(o(T+ΔT))) from switching amplifier904B through a resistor and the input signal (PWM_(di(T))) 903 through aresistor. Similar to embodiment 806, feedback capacitors are connectedbetween the inverted output and the positive input and between thenon-inverted output and the negative input of the difference amplifier907. The inverted and non-inverted outputs are then applied to the loopfilter (−H(s)), which combines them and produces a signal forsample-and-hold (S/H) circuitry that produces the feedback error signal(V_(e)). As discussed with respect to FIG. 8, the feedback error signal(V_(e)) from feedback filter 906 can be provided to the summation blockfor the linear interpolation predictor circuitry within PFE 902.

Thus, embodiment 900 operates to implement a differential BTL switchingamplifier using open loop pulse width adjustment with differential PFCbased on the same PFC principle employed for the single-endedapplication, as discussed above, with a few adjustments. The adjustmentsfor differential PFC 902 from PFC 802 include:

-   -   1. Integration of the varying delta component of the power        supply is weighted by the differential pulse width, created by        XORing the positive (P) and negative (N) input PWM signals 901        and 903 using XOR block 960.    -   2. The complementary threshold voltages for the delay        comparators are cross coupled between the positive (P) pulse and        the negative (N) pulse using MUXs 954 and 956, such that the        P-pulse rising edge will adjust in the opposite direction of the        N-pulse rising edge, and similarly for the respective falling        edges.    -   3. A sign signal (SGN) 922 then controls MUXs 954 and 956 to        determine which direction the respective rising edges adjust in        response to the compensation prediction signal to insure the        compensation is in the correct direction based on the sign of        the signal.

To summarize differential operation of embodiment 900 of FIGS. 9A-B, thesingle-ended pulse widths are adjusted in opposite directions by equalamounts proportional to the differential pulse width with the relativedirection of change determined by the sign signal (SGN) 922.

Closed Loop Pulse Width Adjustment Embodiments (Timing Adjustment)

FIGS. 10A, 10B and 10C are block diagrams for embodiments of predictivefeedback compensation (PFC) circuitry including closed loop pulse widthadjustment to increase robustness of the width adjustment circuitry.FIG. 11 provides a timing diagram associated with the embodiment of FIG.10A. And FIG. 12 provides a more general embodiment for the embodimentsof FIGS. 10A, 10B and 10C.

The open loop PFC technique described above measures errors on thesupply voltage and then adjusts the pulse width of the PWM signal tocorrect for noise on the supply voltage. It is also described above thatthe PFC techniques can measure the output amplitude of the PWM signalused instead of using the supply voltage. While these PFC techniqueshelp to compensate for amplitude errors in the PWM output signal, theamount of cancellation is directly proportional to the accuracy of themeasurement and adjustment. For 60 dB PSR improvement, precision on theorder of 0.0001 is required. Achieving this level of precision requires(1) that the supply absolute and ripple voltage measurement isaccurately made by the PFC circuitry, and (2) that the open loop pulsewidth adjustment is exactly correct. For example, if the noise on thesupply voltage causes the pulse amplitude to be 10% too high, then thepulse width should be decreased by (1−1/1.1) or about 0.0909 times. Toease circuit requirements on the pulse width adjustment, timing feedbackassociated with the pre-compensated PWM signal can be used to set thecorrect amount of pulse width compensation. Feeding back the adjustedpulse width versus the original input pulse width permits residualtiming errors of the pulse width adjustment circuit to be cancelled.

Looking first to FIG. 10A, a closed loop pulse width adjustment circuitembodiment 1000 is depicted that uses uncompensated pulse widthweighting. A PWM input signal (T_(i)) is received by a variable widthblock 1006. The variable width block 1006 also receives a signalrepresenting the total amplitude value (V_(p)) and an error correctionsignal (V_(c)) from the low pass filter (H(z)) 1010 with input(I_(error)). The output of variable width block 1006 is awidth-adjusted, pre-compensated PWM signal (T_(i+c)), whereinT_(i+c)=T_(i)*(1−V_(n)/V_(p)). A fixed delay block 1008 also receivesthe uncompensated PWM input signal (T_(i)) and the signal representingthe total amplitude value (V_(p)). The fixed delay block 1008 thenoutputs the PWM input signal (T_(i)) with a fixed delay to an edgetiming comparator, which also receives the pre-compensated PWM signaloutput by the variable width block 1006. It is noted that the variablewidth block 1006 can be implemented using a rising edge delay cell and afalling edge delay cell, for example, using circuitry based upon theembodiment for an edge delay cell shown in FIG. 5.

In the embodiment depicted, the edge timing comparator includes a risingedge phase (Φ) detector 1012 and a falling edge phase (Φ) detector 1014and four mixers 1020, 1022, 1024, 1026. The rising edge phase (Φ)detector 1012 outputs a first DOWN signal (T_(down)) to mixer 1020 or asecond UP signal (T_(up)) to mixer 1022 that represent rising edgetiming error. Similarly, the falling edge phase (Φ) detector 1014outputs a first DOWN signal (T_(down)) to mixer 1024 or a second signal(T_(up)) to mixer 1026 that represent falling edge timing error. Themixers 1020, 1022, 1024 and 1026 also receive a signal representing theamplitude absolute voltage (V_(p)=(V_(r)+V_(n))) throughvoltage-to-current block (G_(p)) 1004. It is noted that the edge phase(Φ) detectors 1012 and 1014 can be implemented as logic circuitry. It isfurther noted that the mixers 1020, 1022, 1024 and 1026 can beimplemented as charge pump circuits that output a charge based uponUP/DOWN timing signals received from the edge phase (Φ)) detectors 1012and 1014 and based upon the voltages input (V_(p)). The UP/DOWN timingsignals and the voltage input (V_(p)) together determine how much chargeis output in each period by the charge pumps. Conversion to a voltagecontrol signal (V_(c)) then occurs in low pass filter 1010.

Thus, in operation, the rising edge phase (Φ) detector 1012 and thefalling edge phase (Φ) detector 1014 will output its respective DOWNsignal (T_(down)) or UP signal (T_(up)) depending upon the widthadjustment timing relationship between a reference pulse width basedupon the uncompensated PWM input signal (T_(i)) and the pre-compensatedpulse width for the pre-compensated PWM signal (T_(i+c)). In particular,if the time-of-transition for the rising edge for the pre-compensatedPWM signal (T_(i+c)) needs to be reduced (i.e., to occur earlier) tomatch a reference edge provided by the PWM input signal (T_(i)), thenthe rising edge phase (Φ) detector 1012 will output the DOWN signal(T_(down)) to mixer 1020. And if the time-of-transition for the risingedge for the pre-compensated PWM signal (T_(i+c)) needs to be increased(i.e., to occur later) to match a reference edge provided by the PWMinput signal (T_(i)), then the rising edge phase (Φ) detector 1012 willoutput the UP signal (T_(up)) to mixer 1022. Similarly, if thetime-of-transition for the falling edge for the pre-compensated PWMsignal (T_(i+c)) needs to be reduced to match a reference edge providedby the PWM input signal (T_(i)), then the falling edge phase (Φ)detector 1014 will output the DOWN signal (T_(down)) to mixer 1024. Andif the time-of-transition for the falling edge for the pre-compensatedPWM signal (T_(i+c)) needs to be increased to match a reference edgeprovided by the PWM input signal (T_(i)), then the falling edge phase(Φ) detector 1014 will output the UP signal (T_(up)) to mixer 1026. Itis noted the reference edges used by the rising edge phase (Φ) detector1012 can be rising and/or falling edges based upon the PWM input signal(T_(i)), as desired. Similarly, the reference edges used by the fallingedge phase (Φ) detector 1014 can be rising and/or falling edges basedupon the PWM input signal (T_(i)), as desired.

For the embodiment 1000 in FIG. 10A, as indicated above, the mixers1020, 1022, 1024 and 1026 receive a signal representing the amplitudeabsolute voltage (V_(p)=(V_(r)+V_(n))) through voltage-to-current block(G_(p)) 1004. This amplitude absolute or total value (V_(p)) acts toweight the timing error signals. The average output of the edge timingcomparator is the net error output (I_(Y)) of the four mixers,represented by:

I _(Y)=(I _(df) −I _(uf))+(I _(ur) =I _(dr))=(V _(r) +V _(n))G _(p)τ_(c) /T.

-   -   where τ_(c) represents the net difference between the two pulse        widths.

The closed loop pulse width error signal is produced by summation block1016. Summation block 1016 outputs the error correction signal(I_(error)) to low pass filter (H(z)) 1010, which in turn provides thepredictive error correction signal (V_(c)) to the variable width block1006. The summation block 1016 receives the output of the edge timingcomparator, comprised of a falling-edge-up (I_(uf)) signal from mixer1026 as a negative input, a falling-edge-down (I_(df)) signal from mixer1024 as a positive input, a rising-edge-up (I_(u)) signal from mixer1022 as a positive input, and a rising-edge-down (I_(dr)) signal frommixer 1020 as a negative input. In addition, summation block 1016 alsoreceives an input signal (I_(pe)) from amplitude error predictorcircuitry 1002 that is proportional to the amplitude error associatedwith the ripple (or AC) component (V_(n)) of the output pulse amplitude.As depicted, the amplitude error predictor circuitry 1002 includes amixer 1003 that mixes the PWM input signal (T_(i)) with a signalrepresenting the ripple (or AC) component (V_(n)) of the output pulseamplitude through voltage-to-current block (G_(n)) 1001. The amplitudeerror predictor circuitry 1002 then outputs the loop input signal(I_(pe)) to summation block 1016 as a positive input.

The average error signal (I_(error)) produced in the embodiment 1000 canbe represented by the equation:

I _(error) =I _(pe) +I _(Y) =[V _(n) G _(n) τ_(i) /T]+[(V _(r) +V _(n))G_(p) τ_(c) /T],

where V_(n) represents the ripple (or AC) component of the voltagesupply or output pulse amplitude, V_(p)=(V_(r)+V_(n)) represents thesupply or output pulse amplitude absolute voltage, V_(r) represents thedesired or reference output pulse amplitude, τ_(c) represents the PWMpulse width pre-compensation, and τ_(i) represents the input PWM pulsewidth. In steady state, the feedback loop should force I_(error)=0,resulting in a pulse pre-compensation of:

τ_(c)=−τ_(i)(G _(n) /G _(p))(V _(n) /V _(p)),

which is of the desired form for perfect cancellation.

It is further noted that the gain value associated with the edge phasedetection in blocks 1012 and 1014 in units of volt/time can berepresented by the equation:

K _(Φ) =G _(p) *V _(p) /C _(i)

-   -   where C_(i)=I_(error) filter integration capacitor        It is also noted that the gain value associated with the        variable width block 1006 in units of time/volt can be        represented by the equation:

K _(τ) =C _(T)/(G _(d) *V _(p))

-   -   where C_(T)=variable width timing capacitor

FIG. 10B is a block diagram for a closed loop pulse width adjustmentcircuit embodiment 1050 that uses pre-compensated pulse width weighting.The embodiment 1050 in FIG. 10B is similar to embodiment 1000 in FIG.10A in most respects. One difference is in the in the second input tomixers 1020, 1022, 1024 and 1026. Rather than being a signalrepresenting the supply or amplitude absolute voltage(V_(p)=(V_(r)+V_(n))) through voltage-to-current block 1004, mixers1020, 1022, 1024 and 1026 now receive a signal representing the desiredor reference amplitude value (V_(r)=(V_(p)−V_(n))) throughvoltage-to-current block (G_(r)) 1054. Another difference is that themixer 1003 within the amplitude error predictor circuitry 1002 mixes thepre-compensated PWM signal (T_(i+c)) with a signal representing theripple (or AC) component (V_(n)) of the output pulse amplitude throughvoltage-to-current block (G_(n)) 1001. A further difference is that thevariable width block 1006 and the fixed delay block 1008 receive asignal representing the desired or reference output amplitude voltage(V_(r)) rather than the signal representing the supply or amplitudeabsolute voltage (V_(p)). These changes in FIG. 10B adjust therepresentation of the error signal (I_(error)) to be the following:

I _(error) =[V _(n) G _(n)(τ_(i)+τ_(c))/T]+[V _(r) G _(r) τ_(c) /T].

In steady state where the feedback loop forces I_(error)=0, theresulting pulse pre-compensation is given by:

τ_(c)=−τ_(i)(G _(n) /G _(r))[V _(n)/(V _(p) +V _(n)(G _(n) −G _(r))/G_(n))],

which is of the desired form for perfect cancellation if G_(n)=G_(r).

These changes in FIG. 10B also adjust the gain values represented above.In particular, the gain value associated with the edge phase detectionin blocks 1012 and 1014 in units of volt/time is now represented by theequation:

K _(Φ) =G _(r) *V _(r) /C _(i)

and the gain value associated with the variable width block 1006 inunits of time/volt can be represented by the equation:

K _(τ) =C _(T)/(G _(d) *V _(r))

FIG. 10C is a block diagram for a closed loop pulse based widthadjustment circuit embodiment 1070 that uses pre-compensated pulse widthweighting along with a linear interpolation predictor. The embodiment1070 in FIG. 10C is similar to the embodiment 1050 in FIG. 10B in mostrespects. One difference is that the embodiment 1070 modifies thevoltage-to-current block (2G_(b)) 1001 to amplify the ripple (or AC)component (V_(n)) of the amplitude by twice the amount in FIG. 10B.Another difference is that a second delayed path is added from mixer1003 to summation block 1016. In particular, a delay element (½ Z⁻¹)1072 is added between the output of mixer 1003 and the summation block1016. This delay path introduces an additional negative input tosummation block 1016 that represents a delayed version of the amplitudepredictor output signal (I_(ped)). For steady state, the representationof the error signal (I_(error)) defaults to be the same as forembodiment 1050 in FIG. 10B. However, improved performance resultsbecause the amplitude error predictor is more accurate.

FIG. 11 is an example timing diagram 1100 for the closed loop pulsewidth adjustment circuitry of FIG. 10A. As depicted, signal 1102represents an uncompensated PWM input signal (T_(i)) including a pulse1020. Signal line 1104 represents a delayed version (T_(i(delayed))) ofthe uncompensated PWM input signal that has been output by the fixeddelay block 1008 and delayed by a fixed amount of a bias delay (τ₁)1022. The variable width block 1006 outputs a pre-compensated PWM signal(T_(i+c)) using the error correction signal (V_(c)). The rising edgephase detector 1012 and falling edge phase detector 1014 compare thedelayed PWM input signal (T_(i(delayed))) with the pre-compensated PWMsignal (T_(i+c)) to help provide the error correction signal (V_(c)) inoperation with the summation block 1016 and the low pass filter 1010.

Signal line 1106 in FIG. 11 represents a pre-compensated PWM signal(T_(i+c)) that is compared with the a delayed version (T_(i(delayed)))of the uncompensated PWM input signal. As shown in FIG. 11, this phasecomparison produces a rising edge delay (τ_(dr)) 1024 between dottedlines 1112 and 1114 and produces a falling edge delay (τ_(uf)) 1026between dotted lines 1116 and 1118. In particular, the signal line 1108represents the rising-edge-down signal (T_(down(rising))) from risingedge phase (Φ) detector 1012 that includes a pulse 1032 having a widththat provides the rising edge delay (τ_(dr)) 1024. (Signal line 1108represents the DOWN signal (T_(down)) sent to mixer 1020 in FIG. 10A.)The signal line 1110 represents the falling-edge-up signal(T_(up(falling))) from falling edge phase (Φ) detector 1014 thatincludes a pulse 1034 having a width that provides the falling edgedelay (τ_(uf)) 1026. (Signal line 1110 represents the UP signal (T_(up))sent to mixer 1026 in FIG. 10A.) The rising edge delay (τ_(dr)) plus thefalling edge delay (τ_(uf)) represents the total pre-compensation delay(τ_(c)) applied to the uncompensated PWM signal (T_(i)). These delaysare then used to produce the error correction signal (V_(c)) that isapplied to the next pulse for the PWM input signal (T_(i)) to generatethe pre-compensated PWM signal (T_(i+c)).

FIG. 12 is a block diagram for a more general embodiment 1200 for theclosed loop pulse width adjustment embodiments of FIGS. 10A, 10B and10C. The variable width block 1006 receives the PWM input signal (T_(i))and the timing feedback error signal 1204 from the timing comparisoncircuitry 1202 and outputs the pre-compensated PWM signal (T_(i+c)). Thetiming comparison circuitry 1202 generates the timing feedback errorsignal (I_(Y)) 1204 by comparing the pulse width between the PWM inputsignal (T_(i)) and the pre-compensated PWM signal (T_(i+c)) to determinea timing difference and then by weighting this timing difference with anoutput amplitude (V_(p) or V_(r)). For practical implementations, adelay block 1008 may be required to provide a delayed version(T_(i(DELAYED))) of the PWM input signal (T_(i+e)) to the timingcomparison circuitry 1202. The timing feedback error signal (I_(Y)) 1204and the amplitude predictive error correction signal (I_(pe)) from theamplitude error predictor 1002 are provided to the summation/integratorblock 1206. The summation/integrator block 1206 then outputs the errorcorrection signal (V_(c)) to the variable width block 1006. As set forthabove, the timing-based feedback error correction signal (V_(c)) isapplied to the variable width block 1006 in order to set the correctamount of pulse width compensation and to compensate for residual timingerrors in the pre-compensation provided by the variable width block1006.

It is noted that the timing comparison circuitry 1202 correlates to therising and falling edge phase (Φ) detectors 1012 and 1014 and the mixers1020, 1022, 1024 and 1026 in FIGS. 10A-C. The summer/integrator block1206 correlates to the summation block 1016 and the low pass filter1010. And the timing feedback error signal (I_(Y)) 1204 correlates tothe combination of the output signals from mixers 1020, 1022, 1024 and1026. Further, as also shown in FIGS. 10A-C with respect to block 1004and 1054, the gain of the timing comparison circuitry 1202 can beproportional to either an output pulse amplitude total value (V_(p)) oran output pulse amplitude desired value (V_(r)), as desired. And theamplitude error predictor 1002 can receive the PWM input signal (T_(i))or the pre-compensated PWM signal (T_(i+c)), respectively, along withthe ripple or AC component (V_(n)) of the output amplitude. It isfurther noted that the amplitude error predictor 1002 in FIGS. 10A-C and12 correlates to the amplitude error prediction circuitry 204 in FIGS.2A-D, and the other circuitry in FIGS. 10A-C and 12 correlate to thewidth adjustment circuitry 202 in FIGS. 2A-C. In other words, the widthadjustment circuitry in FIG. 12 includes the variable width circuitry1006, the timing comparison circuitry 1202 and the summer/integrator1206, as well as the optional delay block 1008. Advantageously, withrespect to the embodiments in FIGS. 10A-C and 12, by feeding back widthadjustment timing information to the pre-compensation process throughthe use of timing comparison circuitry 1202, closed loop widthadjustment is provided in the system, and residual errors in thepre-compensation process will tend to be canceled out.

Further modifications and alternative embodiments of this invention willbe apparent to those skilled in the art in view of this description. Itwill be recognized, therefore, that the present invention is not limitedby these example arrangements. Accordingly, this description is to beconstrued as illustrative only and is for the purpose of teaching thoseskilled in the art the manner of carrying out the invention. It is to beunderstood that the forms of the invention herein shown and describedare to be taken as the presently preferred embodiments. Various changesmay be made in the implementations and architectures. For example,equivalent elements may be substituted for those illustrated anddescribed herein, and certain features of the invention may be utilizedindependently of the use of other features, all as would be apparent toone skilled in the art after having the benefit of this description ofthe invention.

1. A method for correcting output amplitude errors in switchingamplifiers driven by pulse width modulated (PWM) signals, comprising:receiving a pulse width modulated (PWM) input signal having an inputpulse width; predicting an output pulse amplitude error for the PWMinput signal based on a prior PWM output signal; pre-compensating theinput pulse width for the PWM input signal with a width adjustment basedupon a ratio of the predicted output pulse amplitude error to an outputpulse amplitude weighted by a pulse width; and outputting a PWM outputsignal through a switching amplifier, the PWM output signal having apulse width based upon the pre-compensated pulse width for the PWM inputsignal.
 2. The method of claim 1, wherein the pre-compensating stepcomprises pre-compensating the input pulse width for the PWM inputsignal with a width adjustment based upon a ratio of the predictedoutput pulse amplitude error to an output pulse amplitude total valueweighted by the input pulse width for the PWM input signal to producethe pre-compensated pulse width for the PWM input signal.
 3. The methodof claim 1, wherein the pre-compensating step comprises pre-compensatingthe input pulse width for the PWM input signal with a width adjustmentbased upon a ratio of the predicted output pulse amplitude error to anoutput pulse amplitude desired value weighted by a pre-compensated pulsewidth for a prior PWM input signal to produce the pre-compensated pulsewidth for the PWM input signal.
 4. The method of claim 1, wherein thepre-compensating step comprises adjusting the pulse width by adjustingonly one edge of the PWM input signal.
 5. The method of claim 1, whereinthe pre-compensating step comprises adjusting the pulse width byadjusting both edges of the PWM input signal.
 6. The method of claim 5,wherein the pre-compensating step comprises adjusting both edges of thePWM input signal to provide a symmetric adjustment to the pulse width.7. The method of claim 1, wherein the predicting step comprisesmeasuring a varying or alternating current (AC) component of a supplyvoltage to predict the output pulse amplitude error for the PWM inputsignal based on a prior PWM output signal.
 8. The method of claim 7,wherein the predicting step comprises comparing a total supply voltageto a reference voltage representing a desired output pulse amplitude tomeasure the varying or alternating current (AC) component of the supplyvoltage.
 9. The method of claim 1, wherein the predicting step comprisesmeasuring a varying or alternating current (AC) component of an outputpulse amplitude for the PWM output signal to predict the output pulseamplitude error for the PWM input signal based on a prior PWM outputsignal.
 10. The method of claim 9, wherein the predicting step comprisescomparing an output pulse amplitude total value to a reference voltagerepresenting a desired output pulse amplitude to measure the varying oralternating current (AC) component of the output pulse amplitude for thePWM output signal.
 11. The method of claim 1, wherein the receiving stepcomprises receiving two PWM input signals, such that signal informationresides in a difference between the two signals, wherein the outputtingstep comprises outputting two PWM output signals, and wherein thepre-compensating step comprises adjusting pulse widths for each of thetwo PWM input signals.
 12. The method of claim 1, further comprisingdetermining a feedback error signal by comparing the areas of PWM outputsignals with PWM input signals and utilizing the feedback error signalto further adjust the pulse width for one or more additional PWM inputsignals.
 13. The method of claim 1, further comprising predicting anoutput pulse amplitude error associated with a single prior PWM outputsignal.
 14. The method of claim 1, further comprising predicting anoutput pulse amplitude error associated with a plurality of prior PWMoutput signals.
 15. A digital switching amplifier having output pulseamplitude error correction, comprising: amplitude error predictioncircuitry configured to sense a voltage representing the output pulseamplitude for a PWM output signal, to determine a predicted output pulseamplitude error for a PWM input signal using the sensed voltage, and tooutput a predictive error correction signal proportional to a ratio ofthe predicted output pulse amplitude error to an output pulse amplitudeweighted by a pulse width; width adjustment circuitry coupled to receivethe predictive error correction signal and a PWM input signal having apulse width and configured to output a pre-compensated PWM signal with apre-compensated pulse width representing a width adjustment based uponthe predictive error correction signal; and switching amplifier drivercircuitry configured to receive the pre-compensated PWM signal and todrive a PWM output signal.
 16. The digital switching amplifier of claim15, wherein the predictive error correction signal is based upon a ratioof the predicted output pulse amplitude error to an output pulseamplitude total value weighted by the input pulse width for the PWMinput signal.
 17. The digital switching amplifier of claim 15, whereinthe predictive error correction signal is based upon a ratio of thepredicted output pulse amplitude error to an output pulse amplitudedesired value weighted by a pre-compensated pulse width for a prior PWMinput signal.
 18. The digital switching amplifier of claim 15, whereinthe width adjust circuitry is configured to adjust the pulse width byadjusting only one edge of the PWM input signal.
 19. The digitalswitching amplifier of claim 15, wherein the width adjust circuitry isconfigured to adjust the pulse width by adjusting both edges of the PWMinput signal.
 20. The digital switching amplifier of claim 19, whereinboth edges of the PWM input signal are adjusted to provide a symmetricadjustment to the pulse width.
 21. The digital switching amplifier ofclaim 15, wherein the amplitude error prediction circuitry is configuredto sense a varying or alternating current (AC) component of a supplyvoltage to predict the output pulse amplitude error for the PWM inputsignal based on a prior PWM output signal.
 22. The digital switchingamplifier of claim 15, wherein the amplitude error prediction circuitryis configured to sense a varying or alternating current (AC) componentof an output pulse amplitude for the PWM output signal to predict theoutput pulse amplitude error for the PWM input signal based on a priorPWM output signal.
 23. The digital switching amplifier of claim 15,wherein the PWM input signals comprise two PWM input signals, such thatsignal information resides in a difference between the two signals, thePWM output signals comprise two PWM output signals, and the widthadjustment circuitry is configured to adjust pulse widths for the twoPWM input signals.
 24. The digital switching amplifier of claim 15,wherein the amplitude error prediction circuitry is further configuredto determine a feedback error signal by comparing the areas of PWMoutput signals with PWM input signals and wherein the width adjustmentcircuitry is further configured to utilize the feedback error signal tofurther adjust the pulse width for one or more additional PWM inputsignals.
 25. The digital switching amplifier of claim 15, wherein theamplitude error prediction circuitry is configured to predict an outputpulse amplitude error associated with a single prior PWM output signal.26. The digital switching amplifier of claim 15, wherein the amplitudeerror prediction circuitry is configured to predict an output pulseamplitude error associated with a plurality of prior PWM output signals.